So I've got these 990s....

[NewYorkDave]

Steve, feedback isn't a dirty word around here. This ain't an "audiophile" message board. :green:

PRR, I really doubt that CJ will ever get a chance to lay his dread hacksaw on a McIntosh output. Those transformers are worth a mint (and weigh about as much, too).

In the Mac output transformer, the two primary windings (cathode and plate) are wound bifilar, same number of turns. The primaries and the secondaries are interleaved several times. Two 5.5-pound grain-oriented steel "C" cores enclose the coils in a shell. There's a feedback winding of a few turns, but I'm not sure if it's wound bifilar with the secondary or separately.

Distributing the load between cathodes and plates allows a lower turns ratio, about half that used in conventional push-pull plate output circuits. As an example, a conventional amp using two 6L6s would use a plate-to-plate load impedance of 4K to 6K. The McIntosh used two 1K primary windings, bifilar. The winding technique reduced leakage inductance between the primaries to almost nothing, gave unity coupling, and the lower turns ratio reduced shunt capacitance. The interleaving of primaries and secondaries makes for tight coupling, high efficiency and extended frequency response. I think Steve has already made reference to the role of capacitive coupling between primary and secondary in the extension of high frequency response.

Such a transformer would be digustingly expensive to produce today; it was probably pretty expensive to produce even back then.

 

[SonsOfThunder]

Steve, thanks very much for sharing this info!

If anyone is curious about this:
>>>a company in Oregon who makes very nice PC controlled Distortion Measuring equipment => Audio Precision

 

[PRR]

> Deane always designed the preamps with a fixed Feedback Resistor along with a fixed Feedback capacitor.... A high gain limit resistor (43.2 Ohms...) was chosen and was connected close to the -Input of the opamp to isolate the -Input of the opamp from HF peaking caused by stray capacitance from the -input to ground. This is especially important if the gain controls must be several inches away from the 990. The variable gain resistor was placed between ground and the gain limiting shunt resistor.

Some folks here, you gotta draw 'em a picture:

This has the added advantage that when the pot wiper breaks contact, gain goes to "low", not to "infinite" as in the "Basic" plan.

But it is too late tonight to calculate the resistors.

> At low gains, the signal is so hot that noise is not going to be a problem, even though there might be some academic noise improvement by running the first stage with higher gain.

As a technician in academia: I've done some odd gigs and I could not come up with a session where this theoretical noise-rise would have been a problem, even with better recorders than I have.

> The lowest distortion comes with each amplifier running with the minimum gain needed

But still in that ivory tower, I point out that the first stage (with gains like we usually need) always runs at lower level than the second stage, should have less distortion, and could be run with higher gain without rise of overall THD. While the second stage works at high level and may also be pulling a heavy load; it might be happier with lower gain and more NFB.

But as essayed below, on the 990 the point may be moot:

> lots of feedback (whoops, dirty word?) and very, very low distortion.

People who think feedback is a dirty word don't remember (may not have been around) when most things had little to no feedback. I grew up on naked-pentode table-radios and phonos, and I still shudder to remember that sound. (There were some good ones, but most were wretched.)

But simple no-feedback amps have simple distortion. (I was mildly amazed whan I recently knocked-up a good naked-pentode amp: it is "phatt" but not as nasty as the bad old radios.) When we put 6dB-10dB of NFB on, the THD number goes down, the sound is different, but after a while it grates. Putting output garbage back in the input re-distorts it, and with complex signals (music) lays a broad-spectrum smear of IM products. Low in %, but filling all the cracks between notes.

Some studies show that, for a single FET, and allowing for the low-order THD in the ear, the high-order products that annoy most actually increase at 6dB NFB, then decline slowly, and are not lower than a no-feedback FET until we get around 30dB-40dB NFB.

BJT is different, actually worse, but gain is so much better that big NFB is cheap. Not cheap enough: those naked-pentode phonos were replaced with "hi-fis" that often had only 15db-25dB NFB in each of several stages. Pro gear was often better, but still not enough NFB to really slam the high-order THD down to inaudibility.

So the trend of thought in the late 1970s was to get huge forward gain and thus huge NFB. You got things like the Phase Linear 400, with enormous gain (though falling bad across the audio band). Abuse of this idea revealed slew-rate as one important concern: a low-gain BJT amp will usually slew-limit before its gain falls off. But also many high-gain amps are just short of NFB in the top of the audio band.

The 10mHz gain-bandwidth claimed on Hardy's paper seems "slow". Audiophiles today like 500MHz parts. Doing the math, at 20KHz a 990 only has a forward gain of 500. Fed-back to gain of 100, it only has 5:1 or 14dB of apparent NFB. Yet we know a 990 works well at such gain.

Digging deeper, the 990 has a lot of internal feedback. ~12dB in the input pair, also push-pull helps. The main gain stage has around 24dB NFB. So without any external NFB, the 990 has about 36dB NFB already, enough to suppress high-order products to the edge of inaudibility. 50dB total NFB at 20KHz and gain of 100, a fairly common working point. Still 30dB at gain of 1000, and at that point the gain will be drooping inside the audio band. So the 990 will tend to be very-clean even at huge gains.

It isn't clear that 30dB-40dB effective NFB is "enough", especially in a cascaded BJT system. More ought to be better. Also there is that pesky treble-droop when you work a 990 at very high gain. (If you use a 75:600 output transformer, you can get another 9dB of "free" gain and not have to work the 990 itself at such high gain.)

Doubling-up the amps may double the THD, but it squares the gain-bandwidth product. Two 990s offer an effective GBW of 100THz! That is tempered by 12dB/oct slope. But taken in two steps, each 30dB max, you have about 24dB external NFB plus 36dB internal NFB for an effective 60dB NFB. Which may push high-order THD and IMD products well below the edge of inaudibility. (And offer flat gain to 250KHz, if you like such stuff.)

 

[dpaton]

PRR-

Excellent analysis. Thanks for that. It seems to me that given the inherant NFB inside the 990, there's no really convincing argument either way on the gain staggering, just two ways to get to the same result, with roughly the same performance.

[quote author="PRR"]But it is too late tonight to calculate the resistors. [/quote]
My brain is still recovering from the deluge of NFB info in your post :shock: As a mild aside, are there any merits to the series (like a pot with discreet steps) vs ladder (L-pad constant impedance) schemes for attenuators in this application? It seems the former would be much easier to build than the latter, but would be heavily dependent on getting resistors exactly 3dB apart, which may or may not be easy in the realm of 1K and under (I haven't gotten the math done either). 24 steps of 3dB seems like a good starting point, with a +/-3dB trim pot.

Overall, it seems like the twin-servo is gain-staged very well as-is, and not a lot is to be had by tweaking the internal structure, no?

I can't say I'm a student of preamp design like this, but it seems to me to be very logical, once the obscurity of the NFB of the 990 itself is taken into account.

The idea of flat gain to 250k is very enticing, as I'm quite enamored of my Earthworks mics and the effect that their..ahem...extended response...has on high frequency sources. It seems that Mr Blackmer may have had had something when he said that 20k should not be the limit. At least to my ears it makes a difference. Of course, it'll require transformers that are equally stellar, but I plan to get those anyway.

Steve-

Wonderful post as usual. Can you enlighten me to the siecial features of your pad for the JT-16? Normally, I'd assume it's just a -20dB pad similar to the one shown in Jensen's as016 app note, but after reading your last few replies, I'm convinced there's some special magic working in there. Is it more than just a pair of series resistors and one bridging resistor? Any special ratios to consider?


Samuel-

It looks like your sheet calculates a feedback attenuator, rather than a shunt attenuator (aka variable resistor or series attenuator), as in PRR's schematic. Is that correct?

When I threw in 1K as the value, it gave me >2k as the max attenuation resistor. That didn't seem right?



All, thanks for continuing this wonderful discussion. This is so much better than bashing it out on my own and having to come back twice a day for help. I raise a toast :guinness: to all involved!

Thanks

-dave

 

[Steve Hogan]

To PRR:

Thank you for the drawing. With the values shown I use 560 pF where you had guessed 1000pF?

To Everyone:

There are a couple of very important design features of the 990 that makes its 10 MHz Unity gain frequency a little misleading.

The most important feature was the subject of the 990 patent -- the two inductors in parallel with the emitter degeneration resistors in the differential pair (LM 394). The 30 Ohm emitter resistors are necessary for high frequency stability, but they would ruin the noise performance of the LM 394 Supermatch pair if they were present in the circuit at audio frequencies. By "shorting out" the emitter resistors in the audio band with the parallel inductors, two things happen -- the noise is reduced to very close to the best the LM 394 can do, and the open loop gain of the opamp at low frequencies is boosted 6 dB compared to the circuit with the 30 Ohm resistors in place.

Deane once commented to me that although he claimed the noise improvement in his patent it was his patent attorney who pointed out the gain improvement. This extra open loop gain gives the 990 superb audio capabilities. A careful examination of the open loop gain and phase response plot in the AES paper reveals that the 990 does not have a typical 6 dB per octave slope to its open loop response. The open loop response has kind of a "hump" to it so that it ends up having much higher gain bandwith product in the audio range than its actual 10 MHz unity gain frequency would suggest. The trick with negative feedback is that it must be carefully applied. The internal compensation of the 990 is extraordinarily well thought out to maximize phase margin and stability. Deane and I believe strongly that if an amp has small signal overshoot, the circuit will benefit sonically from eliminating it.

A second feature of the 990 is the way the 2nd stage loads the collector of Q2. The loading here prevents overshoot of the first stage inside the 990. The 990 has no overshoots going on inside it -- it has very low TIM. That is one of the reasons why a 990 is probably the best amplifier around to use as a console summing amp. It doesn't cloud up and flatten out when summing discrete instruments into a complex mix... It has the ability to maintain the individual placement of instruments in the mix because it has such low intermod.

Re: using a step-up output transformer. The fact that the output amp runs at lower gain is pretty much cancelled out by the fact that the amp must drive a significantly lower impedance load, causing inherently more distortion. This is especially true when the output transformer is loaded with 600 Ohms. More importantly, however, is the fact that the transient response of a multifilar output transformer is much worse in step-up mode than a bifilar. The step-up transformers of yesteryear were primarily there to achieve insane headroom specifications. For best sonics in today's circuits which do not need +30dBu output, the 1:1 bifilar transformer will have much better transient response than a step-up.

Re: Proper Pad for JT-16-A/B

The pad topology is exactly as shown in the Jensen application notes but the proper values for the JT-16-A/B are: 735 Ohm series (2 resistors) and 165 Ohm shunt. This is a minimum loss pad of -20.68 dB (close enough to -20 to write that on the panel) This pad has the same input Z as the transformer and presents the transformer with 150 Ohm source which is optimum for best transformer transient response.

Re: Low Z microphone build out resistors:

Don't forget to have a set of switchable resistors in series with the primary in front of the pad to build out the rather low source impedance presented by the current generation of transformerless, low output impedance microphones. After studying the output impedances of dozens of popular mics, I have settled on 60.4 Ohms as the best compromise. The additional series 120.8 Ohms makes 30 Ohm B&K mics look like 150 Ohms and 50 Ohm Neumann mics look like 170 Ohms, which is much better for the transient response of the transformer than lower impedances. Higher Z mics roll off the high end of the input transformer, but mic impedances much lower than 150 Ohms make a peak in the response, which is sonically much, much worse.

 

[PRR]

> With the values shown I use 560 pF where you had guessed 1000pF?

Is that a question or a quote? If quote: I picked a round number that would not shave 20KHz hardly at all; I should have noted that this was a tweak. If someone does not use good layout, even this bandwidth may not be stable; trying 2,000pFd may be quicker than fixing the layout (in DIY). Of course the iron was intended for wide bamdwidth, so 500pFd (0.3MHZ at -3dB!) fits that aim, realizing that 60-some dB of gain at that frequency is an RF problem (it is nearly the gain in a good AM radio IF strip, which has to be lain-out just-so or it will whistle).

> "shorting out" the emitter resistors in the audio band...

Darn darn darn. I knew that; and have mentioned it as one of the brilliant techniques that you won't find in any IC (not just because of patent). But last night I just missed it.

This really is a very subtle circuit. At a glance it is very much like some of the tutorial discrete op-amps on Doug Self's site, but the detailing is superb and makes a big difference in musical use.

> settled on 60.4 Ohms as the best compromise

But, ummm... at first glance, Deane used chokes to take 30+30= 60 ugly ohms out of the emitters, and you put it back in the base?

Ah... "transformerless, low output impedance microphones" really means mikes with amplifiers. And therefore the 990 is not then the lowest-noise point in the chain. And most of these mikes are not only hot enough to overwhelm the noise of a lesser amp, their self-noise is high compared to passive mikes (dynamics and ribbons). My AKG 414s have about 1uV of noise, which makes a 0.2uV or 1dB NF/150Ω noise spec pointless. OTOH my old Beyer ribbons really did like a low-noise amp. And anyway they had enough source impedance to terminate the residual ring that those 30Ω B&K might bring out.

> believe strongly that if an amp has small signal overshoot, the circuit will benefit sonically from eliminating it.

Much the same thing was found objectively in fast-settling analog data systems. You can gimmick-up the phase plot by taking some rolloff in one stage, the rest in another stage. Sort of like fixing a bump with a roll-off. But the slopes of the two stages never mate exactly. The resulting glitch in the phase plot does not hurt stability a bit, even if they are an octave apart. But the settling behavior to high accuracy gets very bad. Where a very objective spec is given, and an an amp that "should" be better won't meet spec, the reason can be found. Audio does not have any single objective spec (we don't have any single signal to capture) so this point is hazy. But controlling overshoot at every stage (not just end-to-end) does seem a valid path.

 

[NewYorkDave]

"shorting out" the emitter resistors in the audio band...

For an earlier example of a similar but not identical concept, check page 545 of Tremaine, second edition. A 5mH inductor is used in series with the cathode of the input tube to cause the stage gain gain to fall at above-band frequencies, with little detriment to the AF gain.

But, ummm... at first glance, Deane used chokes to take 30+30= 60 ugly ohms out of the emitters, and you put it back in the base?

I was wondering about the same thing. Since we're talking about a "switchable" deal either way, I think what I would have wanted to try is to switch in additional termination components on the secondary to tame the rise. But maybe this was already tried and found unsatisfactory compared to simply building out the source impedance...

 

[PRR]

> I think what I would have wanted to try is to switch in additional termination components on the secondary to tame the rise.

Not the same thing.

The leakage inductance decouples the primary and secondary. And the simple lumped-impedance model we like to use is not perfectly accurate. It is possible to have a combination of impedances that would be hard to tame at the secondary, yet tame easily on the primary. It is also possible to have an internal ringing and load the secondary to "flatten" it; in the thought-trend Steve outlined, that would be wrong. A peak plus a droop is not the same as really-flat at every point.

 

[Dan Kennedy]

I would think you could think of it as "Spoiling the Q" of the parasitic effects, and the simplest place is at the input. It's too bad the primaries aren't in series, or at least wired out as so, then it'd be just one resistor to switch.

Hmmm, something to try on a tranny I happen to have...I'll report back.

 

[CJ]

Wow. What a cool discussion. Way over my head.

That isolator resistor does two things. Adds resistance and provides a former for the inductor coil.

Right about the bi-filar/litz type transformers: dc to daylight, no peaking and no phase shift. Somehow they managed to balance the leakeage so the capacitance takes over at the right moment. I did a dissection on the API output as a few might remember.

As far as the McIntosh iron, I have been in touch with the only authorized re-winder of Mac and Marantz transformers for the last few months. His is 73 years old and does alll the winding himself. Obviously he is not going to send me blueprints, but he did tell me that one of his big outputs has 50 primaries! Talk about inter-leaving! And yes, wound on a C-core.

Oh yeah, $2500 bucks a piece and they sell like hot cakes.
He does some of the Fairchild stuff also.

Sorry for the OT.
Carry on!

 

[Samuel Groner]

It looks like your sheet calculates a feedback attenuator, rather than a shunt attenuator (aka variable resistor or series attenuator), as in PRR's schematic. Is that correct?

When I threw in 1K as the value, it gave me >2k as the max attenuation resistor. That didn't seem right?

Did not check PRR's post carefully yet but that's what I came up for your app:
[removed]

Samuel

 

[PRR]

> what I came up for your app:

Looks slick, but there must be an off-by-one error in it some place. The 30dB by 1dB steps values look right, but try 29dB or 28dB: the results look wrong, and become obviously wrong if you pick max gain of 20dB or 60dB.

Ah: many (all?) references to column E should refer column D. That looks right, though I have not tested extensively, nor thought-through the math.

When the range extrapolates to 0dB or less, you get different "errors". Not worth fixing, but need physical explanation. In short, of course, this connection can't give a gain of less than unity (0dB).

0dB gives a DIV/0 error. In reality it needs an infinite shunt resistor, a practical value. You can see this by starting from say 20.001dB: then 0.001dB needs 8,685,389.648 or 8M6. In practice, you can't hear the difference between +0.5dB and 0.0dB: +0.5dB needs a 16876.576 or 17K resistor, which will sound the same as an open circuit. (If there were a capacitor in the feedback, you would want some high-value resistor to keep the cap charged to the offset voltage so it didn't "pop" as you came off of "0dB"; that kinda forces "0dB" to really be like 0.1dB or 0.5dB).

For gains lower than 0dB it computes a negative resistor, which is a "correct" answer, but not a real-world part. (Yeah, we can gyrate a negative resistance, but that is really the hard way to get attenuation.)

 

[Samuel Groner]

many (all?) references to column E should refer column D. That looks right, though I have not tested extensively, nor thought-through the math.

Column D are the theoretical values. They are calculated with reference to the "real" values you entered in column E to correct for differences between calculated values and the exact values you use. To get a reasonable value in i.e. D9, you have to enter a suitable value in E8.

I did put in the values JH has on his schemo for the M-2 and I got close up to 0.5 dB.

But for your pleasure, I will check the math again and try to document this a bit better.

Samuel

 

[John Hardy]

Regarding the schematic in the M-1 and M-2 data package:

http://www.johnhardyco.com/pdf/M1_M2_M1p_20031025.pdf

R7 is listed as 20 ohms but was changed in production to 19.1 ohms long ago to get to a full 60dB total gain (54.4dB of gain from the 990, 5.6dB of gain from the JT-16-B input transformer). I'll update the schematic someday...

Also, when calculating the gain, the resistor that brings the DC servo signal back to the inverting input of the 990 (R15, 200k) must be taken into consideration. It is effectively in parallel with R7 and the other gain-adjust resistors that form the path from the inverting input to ground.

John Hardy
The John Hardy Co.
www.johnhardyco.com

 

[Samuel Groner]

R7 is listed as 20 ohms but was changed in production to 19.1 ohms long ago to get to a full 60dB total gain (54.4dB of gain from the 990, 5.6dB of gain from the JT-16-B input transformer). I'll update the schematic someday..

Ah, that's my 0.5 dB..!

Also, when calculating the gain, the resistor that brings the DC servo signal back to the inverting input of the 990 (R15, 200k) must be taken into consideration. It is effectively in parallel with R7 and the other gain-adjust resistors that form the path from the inverting input to ground.

OK, good point. But probably insignificant with 1k feedback resistance for the Twin Servo.

Samuel

 

[John Hardy]

The DC servos of the Jensen Twin Servo are returned to the non-inverting side of the 990s, so they do not affect the usual gain calculations. However, there is a voltage divider created by R10 and R15 on the Jensen Twin Servo schematic that would cause a loss of a whopping .01dB (or so):

http://www.jensentransformers.com/as/as083.pdf

R3 and R9 are another story. But their effect is also insignificant (more or less).

John Hardy

 

[Samuel Groner]

The DC servos of the Jensen Twin Servo are returned to the non-inverting side of the 990s, so they do not affect the usual gain calculations.

Do you use an inverting servo for your Twin Servo? Intuitively, I would have changed it to a noninverting. Otherwise, there's some DC flowing through the input XFMR secondary, no (purely academic, I know)? The advantage of the inverting servo is that the frequency response does not change with gain, right? Any other advantage?

BTW, why are the servos set to different cutoff points?

Samuel

 

[John Hardy]

The M-1 and M-2 mic preamps (and the MPC-500C, MPC-600 and MPC-3000 mic preamp cards) use a DC servo circuit based on a non-inverting servo that is returned to the inverting input of the 990. The Jensen Twin Servo that I make uses a DC servo circuit based on an inverting servo that is returned to the non-inverting side of the 990. It's the same thing only different.

The Twin Servo that I have been making since 1989 was developed by Deane Jensen and Bill Whitlock, with Steve Hogan having a hand in it too (I think). The DC servos were originally going to be non-inverting, going back to the inverting input of the 990s. But the design was changed late in the development cycle.

My Twin Servo also has input bias current compensation circuits that are not included in the Twin Servo schematic that is shown on the Jensen site. The gain pots on my Twin Servo are strictly in the path from the inverting input to ground. The Jensen app note has the pots partly in the feedback loop.

And before you ask, the schematic for my Twin Servo is not available! Official policy from Jensen!

John Hardy

 

[dpaton]

[quote author="John Hardy"]And before you ask, the schematic for my Twin Servo is not available! Official policy from Jensen![/quote]
I'm honestly sympathetic to the Jensen IP, but if we figure some stuff out based on the lilluminating suggestions from you, Steve, PRR, DK, and the rest, and it happened to end up very close to your version, would ours be fair game for the public? (assuming nothing was misappropriated, just collaborating minds coming to similar conclusions as Deane and Bill and Steve and you did).

I'm curious, becausece I planned on releasing the heavily doctored Group-DIY variant of the JTS schematic back to the group here once I get all the suggestions and changes comitted. I don't want to step on anyone's toes, but it'd be a shame if all this sharing couldn't be compiled in a nice little package.

All- I haven't posted much 'cause I'm watching and reading just like the rest of you. The schematic is still in process, and I'll post an update in a week or so when things slow down at work, assuming no one minds :roll:

-dave

 

[Guest]

I've been following this too, as I have a twin servo in the books... Still trying to get my head around a little bit of this... Very enlightening, and much appreciated comments/explanations.